Processing of Multi-Carrier Signals Before Power Amplifier Amplification

ABSTRACT

Embodiments for methods and apparatuses for processing a multi-carrier signal are disclosed. One method includes shaping a frequency spectrum of a multi-carrier transmit signal wherein an amplitude of a plurality of subcarriers of the multi-carrier transmit signal is increased relative to at least one other subcarrier of the multi-carrier transmit signal. The shaped frequency spectrum multi-carrier transmit signal is amplified with a power amplifier, wherein a power level of an output of the power amplifier is greater than a rated power level of the power amplifier.

RELATED APPLICATIONS

This patent application claims priority to U.S. provisional patentapplication Ser. No. 61/209,902 filed on Mar. 11, 2009 which isincorporated by reference.

FIELD OF THE DESCRIBED EMBODIMENTS

The described embodiments relate generally to wireless communications.More particularly, the described embodiments relate to preprocessing ofmulticarrier signals before amplification by a power amplifier.

BACKGROUND

Conventional wireless systems employ radio-frequency (RF) transmittersto produce an output signal that can be applied to an antenna forcommunication between stations separated by some distance. In mobilewireless networks, one station may be a subscriber station (SS), whereasanother station may be a base station (BS). As the SS roams throughoutthe coverage area of the wireless network, the path loss between the SSand the BS changes due to a number of factors including the change indistance between the stations as well as the presence of objects in theenvironment that serve to obstruct or attenuate the signals travelingfrom one station to the other. To ensure proper network operation, theBS will instruct the SS to increase or decrease its transmit power asrequired to overcome the path loss between the SS and BS so that the BSwill continue to receive the MS signals as channel conditions change.Over the full range of possible transmit powers, the SS must maintain acertain signal quality so as not to inhibit detection of its transmitsignals by the BS. Depending upon the details of the physicalenvironment between the SS and BS, at some critical distance from the BSthe SS will no longer be able to increase its output power whilemaintaining the required signal quality. At that point, communicationbetween the SS and BS can no longer be maintained and the link will bedropped unless the BS is able to hand-off communication with the SS to aneighboring BS. Therefore, the maximum output power capability of the SSis: a critical parameter that ultimately determines the expecteddistance over which the SS and BS can communicate and thereby the numberand spacing of BS sites that is required to provide reliable coverage ina mobile network. However, the greater the number of BS sites, thegreater the cost to implement the mobile network.

Accordingly, there is a need to maximize the output power capability ofthe MS to ensure reliable coverage with a minimum of required BS sites.The coverage is usually limited by the MS as the BS transmittertypically has sufficient output power to provide reliable coverage overan acceptable cell area.

It is instructive to consider the factors limiting the maximumtransmitter output power in a conventional RF transmitter. Among thosefactors are the error vector magnitude (EVM) and the spectral emissionsmask. The EVM characterizes the fidelity of the actual transmit signalwith respect to the intended transmit signal. This is commonlyvisualized as illustrated in FIG. 1 in which the complex transmittedsignal comprising in-phase (I) and quadrature (Q) components at certaincritical instants in time is compared to a regular constellation ofpoints representing the ideal values of the transmitted signal at thosesame instants. The constellation of points that are used in transmissionis referred to as the modulation. The EVM is given by theroot-mean-square (RMS) distance between the actual signal and thecorresponding ideal constellation points normalized to the averageradius over all of the constellation points. Forward error correctioncodes are commonly used in wireless transmission. Taking together, themodulation and the coding schemes are referred to as the Modulation andCoding Scheme (MCS). Different Modulation and Coding Schemes havedifferent EVM requirements. A greater EVM can be tolerated for a‘loosely packed’ constellation corresponding than it can for a ‘denselypacked’ constellation corresponding for the same coding rate. In manysystems, the transmitter may be able to operate using a variety of MCSlevels. Doing so allows for the transmission data rate to be adapted asconditions allow. For example, when the MS is closer to the BS, the BSwill generally be able to detect a higher MCS level thereby allowing foran increased data rate for data transmitted from MS to BS. Similarly,when the MS is farther from the BS, the BS may need to reduce the MCSlevel to ensure reliable reception. Thus, having some flexibility tocontrol the MCS level is advantageous in that it provides the ability tooperate at the maximum data rate that can be accommodated by the linkconditions. The transmitter EVM is degraded by noise and intermodulationdistortion products produced by the transmitter as it amplifies thetransmit signal.

A second factor limiting the maximum transmitter output power is thespectral emissions mask, which characterizes the amount of spuriousemissions generated by the transmitter that fall into neighboringchannels. As illustrated in FIG. 2, there is a limit on the acceptablelevel of such emissions to avoid interference with neighboringtransmitters. These emissions are caused primarily by inter modulationdistortion of the, transmit signal occurring due to nonlinearamplification by the transmitter. Hence, both EVM and spectral emissionsmask performance are determined by noise and nonlinearity in the RFtransmitter.

A critical component in a conventional transmitter that produces suchdistortion is a power amplifier. A power amplifier will typicallypossess a maximum output power rating. Operating the power amplifier atoutput powers exceeding this rating may result in unacceptable EVM orspectral mask performance. As an RF transmitter may be asked to producethe maximum output power for any MCS level, it is generally necessaryfor the transmitter to comply with the most restrictive EVM requirementcorresponding to the highest MCS level while also meeting the spectralemissions mask.

However, when the MS is positioned near the outer boundary of a given BScell, the RF transmitter may be operating at a lower MCS level because alower MCS level is more tolerant of attenuation along the path betweenMS and BS and therefore is easier to detect and demodulate. Under suchoperating conditions, one can infer based on the foregoing discussionthat the maximum output power of the transmitter is primarily dictatedby the spectral emissions mask requirement rather than the EVMrequirement since the latter enjoys a relaxation for low MCS levels.However, a relaxed EVM requirement alone is not enough to permitoperation of the transmitter at an increased output power because thetransmitter must satisfy the tighter specification imposed by thespectral emissions mask requirement which is typically independent ofMCS level.

It is desirable to have a technique that allows for increased outputpower at low MCS levels at the expense of EVM performance whilemaintaining a specified spectral emissions mask performance. Doing sowould enable a beneficial increase in transmitter output power when theMS operates near its maximum range from the BS, thereby improving thereliability of the network and reducing the required number and spacingof base stations. An object of the present invention is to provide thiscapability.

SUMMARY

An embodiment includes a method of processing a multi-carrier signal. Afirst step of the method includes shaping a frequency spectrum of amulti-carrier transmit signal wherein an amplitude of a plurality ofsubcarriers of the multi-carrier transmit signal is increased relativeto at least one other subcarrier of the multi-carrier transmit signal. Asecond step of the method includes amplifying the shaped frequencyspectrum multi-carrier transmit signal with a power amplifier, wherein apower level of an output of the power amplifier is greater than a ratedpower level of the power amplifier.

Another embodiment includes another method of processing a multi-carriersignal. The method includes amplitude compressing a time-domain versionof the multi-carrier transmit signal and filtering the compressedmulti-carrier transmit signal. The compressed multi-carrier transmitsignal is amplified with a power amplifier, wherein a power level of anoutput multi-carrier signal of the power amplifier is greater than arated power level of the power amplifier.

Other aspects and advantages of the described embodiments will becomeapparent from the following detailed description, taken in conjunctionwith the accompanying drawings, illustrating by way of example theprinciples of the described embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows an example of an I-Q modulation constellation showing anexample of an EVM.

FIG. 2 shows an example of a frequency spectrum of an OFDM signal, and atarget spectral mask.

FIG. 3 shows an example of an OFDM transmitter that includes time-domainnonlinear amplitude compression.

FIG. 4 shows an example of representative compressive nonlinearitiesconsisting of p-norms and a polyhedral norm.

FIG. 5 shows another example of an OFDM transmitter that includestime-domain processing in which the real and imaginary components of thetransmit signal are individually compressed.

FIG. 6 shows an example of compressive nonlinearity, wherein a phaseangle is preserved.

FIG. 7 shows an example of CORDIC function processing that can be usedto amplitude compress the time domain transmit signal.

FIG. 8 is an example of a block diagram of a transmitter that includesspectral shaping in the frequency domain.

FIG. 9 is an example of a block diagram of a transmitter that includesspectral shaping in the time domain using digital signal processingtechniques.

FIG. 10 is an example of a block diagram of a transmitter that includesspectral shaping and a compressive nonlinearity.

FIG. 11 is a flow chart that includes an example of a method ofprocessing a multi-carrier signal.

FIG. 12 shows an example of representative spectral shaping functions.

FIG. 13 is a flow chart that includes steps of another example of amethod of processing a multi-carrier signal.

DETAILED DESCRIPTION

The embodiments described include methods and apparatuses for increasingthe output power of an OFDM (Orthogonal Frequency Division Multiplexing)RF (Radio Frequency) transmitter. OFDM transmitter processing of thetransmit signal can be include several steps that may be applied.individually or in combination to allow for increased output power atthe expense of EVM performance while maintaining spectral emissions maskperformance. For an embodiment, the processing steps include a frequencyshaping step that tailors the frequency response of the signal. For anembodiment, the transmit signal is compressed by a memory-lessnonlinearity to produce a compressed signal. The compressed signal isfiltered to produce a filtered compressed This signal is then coupled toan RF power amplifier. For another embodiment, the processing stepsinclude a frequency shaping step that tailors the frequency response ofthe signal followed by compression by a memoryless-nonlinearity. Thecompressed shaped signal is filtered to produce a filtered compressedshaped signal. This signal is then coupled to an RF power amplifier.

FIG. 3 shows an example of an OFDM transmitter that includes time-domainnonlinear amplitude compression. The OFDM transmitter can beimplemented, for example, in a subscriber station (SS). Symbol data(310) is applied to the input of circuit which implements an inverseFast-Fourier transform (IFFT) (311). The real and imaginary componentsof the IFFT outputs are upsampled by upsamplers 312 a and 312 b, whichinterdigitate zeros between the samples of the IFFT output. The outputsof upsamplers 312 a and 312 b are applied to digital filters 314 a and314 b, which filter the real and imaginary component of the basebandsignal. For this embodiment, the outputs of the digital filters arecoupled to a memoryless compressive nonlinearity function 315. Thecompressive nonlinearity is a nonlinear circuit element which isapproximately linear for small signal inputs and has reduced gain forlarger signal inputs. Let x=x_(r)+jx_(i)εC² denote the input to thecompressive nonlinearity, where and x, denote the real and imaginarycomponents of x, respectively. Define

x = [ x r x i ] ∈ 2

denote the direct sum representations of the real and imaginarycomponents of x. Let f(•:

→

denote the compressive nonlinearity. Then, ∥f(ax)∥≦α∥f(x)∥ for α≧1, αε

, for the appropriate choice of norm.

The compressive nonlinearity is used to limit the peak to average powerratio of the OFDM signal at the power amplifier. When a power amplifieris driven to a point where it distorts, the distortion products maycause the spectral mask to be violated. As OFDM is the sum of a numberof sinusoids, it exhibits a large peak to average power ratio (PAPR). Itis the peaks of the OFDM signal which generally limit the maskcompliance of a PA when driven by an OFDM signal. Hence, by limiting thePAPR using a compressive nonlinearity, the output power of the PA may beincreased without violating the spectral mask. This allows the PA toproduce more power than its rated power.

The outputs of the memoryless compressive nonlinearity function areapplied to digital-to-analog converters (DAC) 316 a and 316 b, theoutputs of which is filtered by analog filter 318 a and 318 b,respectively. These analog filters attenuate the replicas of thespectrum which appear at harmonics of the DAC sample frequency. Theoutputs of analog filter 318 a and 318 b, which correspond to thein-phase and quadrature components of the transmit signal, are appliedto an RF upconverter 320. RF upconverter 320 translates the frequency ofthe baseband signal to the desired transmit frequency. The output of theRF upconverter is amplified by power amplifier (PA) 322 and applied toantenna 324.

FIG. 4 shows an example of representative compressive nonlinearitiesconsisting of p-norms and a polyhedral noun of a vector comprising twocomponents. These components correspond to the teal and imaginarycomponents of the transmitted signal. The transfer characteristics ofinclude 4 compressive nonlinearities known as clippers in which thevarious norms of the input signal are limited to unity. Three of thesecharacteristics correspond to c norm limits on the input signal. Thel_(p) of a two element vector is defined as

∥ x ∥_(p)=(|x _(r)|⁹ +|x _(i)|^(p))^(1/p).

It has been determined experimentally the clipping the l₂ norm, ormodulus, of the OFDM signal works well in practice as a compressivenonlinearity. In this case, the signal may profitably be clipped at avalue of 10 dB above the RMS value of the OFDM signal. For transmissionswith fewer constellation points, e.g., QPSK, the clipping may be appliedat value that is lower than 10 dB; this allows transmission of morepower without violation of relevant spectral masks. Specifically, thelevel of compression may be profitably adapted given the desired outputpower and the MCS of the signal to be transmitted.

The octagonal shaped clipper shown in FIG. 4 corresponds to a polyhedralnorm and is denoted by the bold dashed line. A polyhedral norm can bedefined for our purposes as:

∥ x ∥_(poly) =sup _(i=1) ^(n) c _(i) x−d _(i)

where n is the number of functions used in defining the norm, sup refersto supremum, c_(i)ε

² is a row vector, d_(i)ε

, and

denotes the set of real numbers. A polyhedral norm may be implementedefficiently and can be used to implement l₁ norm. A polyhedral noun canalso be used to approximate the l₂ norm.

FIG. 5 shows another example of an OFDM transmitter that includestime-domain processing in which the real and imaginary components of thetransmit signal are individually compressed. This corresponds tolimiting the infinity norm of the input signal. The approach ofindividually compressing the real and imaginary components of thetransmit signal has the advantage of a simple implementation, althoughit its performance is somewhat worse than that of compressivenonlinearities that are responsive to both the real and imaginarycomponents of the transmit signal in general and, specifically, phasepreserving compressive nonlinearities.

FIG. 6 shows an example of compressive nonlinearity, wherein a phaseangle is preserved. Let x_(in)=I_(in)+jQ_(in) denote the input to thecompressive nonlinearity, where I_(in)ε

and a Q_(in)ε

denote the real and imaginary components of x_(in), respectively.Similarly, let x_(out)=I_(out)+jQ_(out) denote the output of thecompressive nonlinearity, where I_(out)ε

and Q_(out)ε

denote the real and imaginary components of x_(out). The compressivenonlinearity preserves phase if the angle of the output of thecompressive nonlinearity x_(out), denoted as φ, equals the angle ofinput to the compressive nonlinearity x_(in).

FIG. 7 shows an example of block diagram of CORDIC function processingthat can be used to amplitude compress the time domain transmit signalin a manner that preserves the angle of the input signal. The termCORDIC (COordinate Rotation DIgital Computer) refers to a method ofperforming trigonometric and other functions without a hardwaremultiplier. CORDIC implementations of functions can be implementedefficiently in hardware. This example includes a CORDIC basedcompression circuit that amplitude compresses, while preserving theangle of a complex signal.

The first step of the CORDIC compression circuit is to calculate|I_(in)| and |Q_(in)| so that further operations can be done in thefirst quadrant of the complex plane. Here |•| denotes absolute value.

The CORDIC compression circuit further includes M forward CORDIC stepsand M inverse CORDIC steps The forward CORDIC step is given by:

I _(f)(n+1)=I _(f)(n)+2^(−n) Q _(f)(n)sign(Q _(f)(n))

Q _(f)(n+1)=−2^(−n) I _(f)(n)sign(Q _(f)(n))+Q _(f)(n).

Here, n denotes the index of the CORDIC recursion, I_(f)(n) and Q_(f)(n)denote the input real and imaginary components of the input to therecursion; I_(f)(n+1) and Q_(f)(n+1) denote the outputs. FIG. 7 showsfour (M=4) such CORDIC steps, with n=0, 1, 2, 3, respectively. For largevalues of M, the Q component output of the final CORDIC recursion isapproximately zero, and hence can be neglected. The corresponding I ourt represents a the magnitude of the input complex signal multiplied by ascale factor

${K = {\prod\limits_{i - 1}^{M}\; \sqrt{1 + 2^{{- 2}\; t}}}},$

this scale factor represents the growth associated with the CORDICoperations. For large M K≈1.6468.

The output of the M th forward CORDIC step, I_(f)(M) is compressed usinga non linear function to produce an intermediate signal I_(r)(M). Onespecial case of interest is the clipping function, defined as,

${I_{r}(M)} = \left\{ {\begin{matrix}\frac{I_{f}(M)}{K^{2}} & {{I_{f}(M)} < {\overset{\_}{\rho}K}} \\\frac{\overset{\_}{\rho}}{K} & {{I_{f}(M)} \geq {\overset{\_}{\rho}K}}\end{matrix},} \right.$

where ρ denotes the modulus of the output in the presence of largeinputs. For inputs whose modulus is less ρ, the output of thecompressive nonlinearity equals the input. To save power, if the resultof the forward CORDIC operations determines that the input signal doesnot require clipping, the input to the first stage of the COMIC isoutput as a result. This avoids the need to compute the quantity

$\frac{I_{f}(M)}{K^{2}}.$

The reverse CORDIC operations are executed if the input signal requiresclipping as indicated by the signal C_(enable), where

$C_{enable} = \left\{ {\begin{matrix}0 & {{I_{f}(M)} < {\overset{\_}{\rho}K}} \\1 & {{I_{c}(M)} \geq {\overset{\_}{\rho}K}}\end{matrix}.} \right.$

Each inverse CORDIC step is defined as:

I _(r)(n)=I _(r)(n+1)−2^(−n) Q _(r)(n+1)sign(Q _(f)(n))

Q _(r)(n)==2^(−n) I _(r)(n)sign(Q _(f)(n))+(Q _(r)(n))+Q _(r)(n).

The final output of the CORDIC compression circuit is defined as:

I _(mux) =I _(r)(0)sign(I _(in))

Q _(mux) =Q _(r)(0)sign(Q _(in))

The C_(enable) signal is used to multiplex the complex signal comprisingI_(mux) and Q_(mux) with the signal input comprising I_(in) and Q_(in),according to

$I_{out} = \left\{ {{\begin{matrix}I_{in} & {C_{enable} = 0} \\I_{mux} & {C_{enable} = 1}\end{matrix}\mspace{14mu} {and}\mspace{14mu} Q_{out}} = \left\{ {\begin{matrix}Q_{in} & {C_{enable} = 0} \\Q_{mux} & {C_{enable} = 1}\end{matrix}.} \right.} \right.$

FIG. 8 is an example of a block diagram of a transmitter that includesspectral shaping in the frequency domain. This embodiment includes thespectral shaping being performed by multiplying the frequency domainrepresentation of the transmitted signal by a windowing function 810.

FIG. 9 is an example of a block diagram of a transmitter that includesspectral shaping in the time domain. Here, the shaping filter isimplemented in the time domain. In another embodiment, the shapingfilter functionality may be included in the digital filter. In yetanother embodiment, the shaping filter functionality may be included inthe analog filter.

FIG. 10 is an example of a block diagram of a transmitter that includesspectral shaping and a compressive nonlinearity. For this embodiment,the use of a compressive nonlinearity is combined with spectral shaping.

FIG. 11 is a flow chart that includes an example of a method ofprocessing a multi-carrier signal. A first step 1110 includes shaping afrequency spectrum of a multi-carrier transmit signal wherein anamplitude of a plurality of subcarriers of the multi-carrier transmitsignal is increased relative to at least one other subcarrier of themulti-carrier transmit signal. A second step 1120 includes amplifyingthe shaped frequency spectrum multi-carrier transmit signal with a poweramplifier, wherein a power level of an output of the power amplifier isgreater than a rated power level of the power amplifier.

For an embodiment, shaping the frequency spectrum of the multi-carriertransmit signal includes increasing an amplitude of a first plurality ofsubcarriers relative to a second plurality of subcarriers, wherein thefirst plurality of subcarriers occupy frequencies that are closer to acenter frequency of the multicarrier signal than the second plurality ofsubcarriers. That is, the frequency offset between the between the firstplurality of subcarriers and the center frequency of the multicarriersignal is smaller (less) that a frequency offset between the secondplurality of subcarriers and the center frequency of the multicarriersignal. For a baseband signal, the center frequency can be zero.However, for an intermediate frequency (IF) or radio frequency (RF), thecenter frequency has a value.

For the described embodiments, the rated power level of the poweramplifier is a maximum power of an output signal of the power amplifierthat meets EVM and spectral mask limits for a standard compliantmulticarrier transmit signal, wherein the standard compliantmulticarrier transmit signal has not been subject to frequency spectrumshaping. A standard compliant multicarrier transmit signal can bedefined by a wireless standard such as WiMAX (Worldwide Interoperabilityfor Microwave Access) or LTE (Long Term Evolution).

For an embodiment, shaping the frequency spectrum of a multi-carriertransmit signal results in the amplified multicarrier signal complyingwith a spectral mask. For another embodiment, the amplified multicarriersignal does not exceed predetermined spectral mask limits. For anotherembodiment, the amplified multicarrier signal does not exceed apredetermined EVM limit.

An embodiment further includes amplitude compressing a time-domainversion of the multi-carrier transmit signal, and filtering thecompressed shaped frequency spectrum multi-carrier transmit signal priorto amplifying the shaped frequency spectrum multi-carrier transmitsignal with a power amplifier. For a more specific embodiment, amplitudecompressing the time-domain version of the multi-carrier transmit signalis responsive to in-phase (I) and quadrature-phase (Q) components of thetime-domain version of the multi-carrier transmit. For an even morespecific embodiment, amplitude compressing the time-domain version ofthe multi-carrier transmit signal comprises processing I and Qcomponents of the time-domain version of the multi-carrier transmitsignal utilizing a plurality of CORDIC operations.

FIG. 12 shows an example of representative spectral shaping functions. Afirst shaping function corresponds to a constant value over the range ofsubcarriers from

$\left\lbrack {{- \frac{N_{used}}{4}},\frac{N_{used}}{4}} \right\rbrack$

with raised cosine responses in the intervals

$\left\lbrack {{- \frac{N_{used}}{2}},\frac{N_{used}}{4}} \right)\mspace{14mu} {and}\mspace{14mu} {\left( {\frac{N_{used}}{4},\frac{N_{used}}{2}} \right\rbrack.}$

In a second spectral shaping function, a trapezoidal shaping function isused. It similarly has a constant value over the range of subcarriersfrom

$\left\lbrack {{- \frac{N_{used}}{4}},\frac{N_{used}}{4}} \right\rbrack.$

It decays at a rate of 0.05 dB per subcarrier to a minimum value. It isdesirable to limit the change in amplitude per subcarrier for tworeasons. First, abrupt changes in amplitude increase the apparent delayspread of the receive signal. Second, some OFDM systems, such as WiMAXhave Radio Conformance Tests with regulate the difference in transmittedpower between adjacent subcarriers. WiMAX is defined in the IEEEstandard P802.16Rev2/D1 (October 2007) and subsequent revisions of thestandard.

Windowing decreases the power transmitted on carriers near the bandedge. This Causes reduced performance on those subcarriers. However, theaggregate effect of boosting the center subcarriers and attenuating theones near the hand edges is still positive.

$\overset{\_}{C} = {\frac{1}{N}{\sum\limits_{k = 1}^{N}{\log_{2}\left( {1 + \frac{{PW}_{k}}{No}} \right)}}}$

An equivalent signal to noise ratio for the collection of subcarrierscan be calculated using

SNR_(EQ)=2 ^(C) −1

If the used subcarriers are approximately uniformly distributed over theinterval

$\left\lbrack {{- \frac{N_{used}}{2}},\frac{N_{used}}{2}} \right\rbrack,$

the effect of the windowing on capacity is minimal and the gains inequivalent SNR are approximately equal to the increase in transmittedpower. The window function may be optimized to according to a predefinedmetric such as the equivalent signal to noise ratio subject to meetingthe spectral mask.

Another embodiment includes a method of processing a multi-carriersignal of a mobile subscriber prior to the multi-carrier signal beingamplified by a power amplifier of the mobile subscriber. A first stepincludes shaping a frequency spectrum of a multi-carrier transmit signalwherein an amplitude of a plurality of subcarriers of the multi-carriertransmit signal is increased relative to at least one other subcarrierof the multi-carrier transmit signal. A second step includes amplifyingthe shaped frequency spectrum multi-carrier transmit signal with a poweramplifier, wherein a power level of an output of the power amplifier isgreater than a rated power level of the power amplifier. For anembodiment, shaping the frequency spectrum of the multi-carrier transmitsignal includes increasing an amplitude of a first plurality ofsubcarriers relative to a second plurality of subcarriers, wherein thefirst plurality of subcarriers occupies frequencies that are closer to acenter frequency of the multicarrier signal than the second plurality ofsubcarriers.

Various embodiments include initiating the frequency spectrum shapingand operation of the power amplifier above its rated power level basedon activities of the subscriber. That is, embodiments includeselectively utilizing the frequency spectrum shaping and operation ofthe power amplifier above its rated power level. An embodiment includesthe frequency spectrum shaping and operation of the power amplifierabove its rated power level being utilized during a wireless networkentry procedure of the subscriber. Another embodiment includes thefrequency spectrum shaping and operation of the power amplifier aboveits rated power level being utilized when the subscriber is handing offfrom a first wireless base station to a second wireless base station.Another embodiment includes the frequency spectrum shaping and operationof the power amplifier above its rated power level being utilized for asubset of the MCS levels available for transmission by the subscriber.Another embodiment includes the frequency spectrum shaping and operationof the power amplifier above its rated power level being utilized for asubset of the transmission modes defined by a standard. For example, thefrequency shaping and operation of the power amplifier above its ratedpower level may be used when in a WiMAX Band Adaptive Modulation andCoding (BAMC) mode but not when in a Partial Usage of Subchannels (PUSC)mode.

As described, an embodiment includes the subscriber station (SS)selecting to use the shaping of the frequency spectrum and transmittinga multicarrier signal at a power level that exceeds the rated power ofthe power amplifier during the network entry process. The network entryprocess is one in which the SS informs the BS of its capabilities andregisters on the network. In some wireless systems, HARQ (HybridAutomatic Repeat Request) is not supported during all stages of networkentry; hence, the additional power output can be used to improve uplinkcoverage.

Another embodiment includes selective use of shaping of the frequencyspectrum and transmitting a multicarrier signal at a power level thatexceeds the rated power of the power amplifier during at least one HARQretransmissions. The link performance of the subscriber may be improvedby increasing the SS transmitter power spectral density during HARQretransmissions. If a sufficient number of HARQ retransmissions do notresult in effort free decoding of the SS transmission, the latency of SSdata may increase disproportionately.

Another embodiment uses shaping of the frequency spectrum andtransmitting a multicarrier signal at a power level that exceeds therated power of the power amplifier for a subset of modulation schemes.An example of this embodiment would be the use of shaping of thefrequency spectrum and transmitting a multicarrier signal at a powerlevel that exceeds the rated power of the power amplifier only whentransmitting QPSK.

Another embodiment uses compressing the time domain version of themulticarrier transmit signal and transmitting a multicarrier signal at apower level that exceeds the rated power of the power amplifier duringnetwork entry.

Another embodiment uses compressing the time domain version of themulticarrier transmit signal and transmitting a multicarrier signal at apower level that exceeds the rated power of the power amplifier duringHARQ retransmissions.

Another embodiment uses compressing the time domain version of themulticarrier transmit signal and transmitting a multicarrier signal at apower level that exceeds the rated power of the power amplifier for asubset of modulation schemes.

Another embodiments uses frequency shaping and compressing the timedomain version of the multicarrier transmit signal when transmitting amulticarrier signal at a power level that exceeds the rated power of thepower amplifier for a subset of modulation schemes.

Another embodiments uses frequency shaping and compressing the timedomain version of the multicarrier transmit signal when transmitting amulticarrier signal at a power level that exceeds the rated power of thepower amplifier during network entry.

Another embodiments uses frequency shaping and compressing the timedomain version of the multicarrier transmit signal when transmitting amulticarrier signal at a power level that exceeds the rated power of thepower amplifier during HARQ retransmissions.

FIG. 13 is a flow chart that includes steps of another example of amethod of processing a multi-carrier signal. A first step 1310 includesamplitude compressing a time-domain version of the multi-carriertransmit signal. A second step 1320 includes filtering the compressedmulti-carrier transmit signal. A third step 1330 includes amplifying thecompressed multi-carrier transmit signal with a power amplifier, whereina power level of an output multi-carrier signal of the power amplifieris greater than a rated power level of the power amplifier.

For an embodiment, amplitude compressing enables compliance with aspectral mask as measured at an output of the amplifier. For anembodiment, the compression of the time-domain version of themulti-carrier transmit signal increases with an amplitude of themulti-carrier signal. For an embodiment, amplitude compressing thetime-domain version of the multi-carrier transmit signal is responsiveto in-phase (I) and quadrature-phase (Q) components of the time-domainversion of the multi-carrier transmit. For a specific embodiment,amplitude compressing the time-domain version of the multi-carriertransmit signal preserves (or at least substantially preserves) an angleof I and Q components of the time-domain version of the multi-carriertransmit.

For a more specific embodiment, amplitude compressing time-domainversion of the multi-carrier transmit signal includes processing I and Qcomponents of the time-domain version of the multi-carrier transmitsignal utilizing a plurality of CORDIC operations. For an even morespecific embodiment, compressing the time-domain version of themulti-carrier transmit signal further includes selecting between theprocessed I and Q components of the time-domain version of themulti-carrier transmit signal and the I and Q components of thetime-domain version of the multi-carrier transmit signal. The input tothe CORDIC can be selected when the input (I, Q) signal has a smallmodulus.

More generally, for an embodiment, amplitude compressing the time-domainversion of the multi-carrier transmit signal includes applying amemory-less compressive nonlinearity function to the time-domain versionof the multi-carrier transmit signal. For a more specific embodiment,the memory-less compressive nonlinearity function limits a l_(p) of themulti-carrier transmit signal. For another more specific embodiment, thememory-less compressive nonlinearity function is a polyhedral norm. Foranother more specific embodiment, the memory-less compressivenonlinearity function operates on a modulus of the time-domain versionof the multi-carrier transmit signal.

Although specific embodiments have been described and illustrated, theembodiments are not to be limited to the specific forms or arrangementsof parts so described and illustrated.

1.-30. (canceled)
 31. A transmitter, comprising: a first digitalprocessing branch configured to process an in-phase component of adigital input signal to provide a processed in-phase signal; a seconddigital processing branch configured to process a quadrature phasecomponent of the digital input signal to provide a processed quadraturephase signal; a compressive nonlinearity module configured to amplitudecompress the processed in-phase signal and the processed quadraturephase signal to provide an amplitude compressed in-phase signal and anamplitude compressed quadrature phase signal, respectively, a firstanalog processing branch configured to process the amplitude compressedin-phase signal to provide a processed amplitude compressed in-phasesignal; a second analog processing branch configured to process theamplitude compressed quadrature phase signal to provide a processedamplitude compressed quadrature phase signal; a upconverter moduleconfigured to frequency translate the amplitude compressed in-phasesignal and the amplitude compressed quadrature phase signal to provide afrequency translated baseband signal; and an amplifier, having a ratedpower level, configured to amplify the frequency translated basebandsignal to a power level that is greater than the rated power level. 32.The transmitter of claim 31, wherein the first digital processing branchcomprises: a first upsampler configured to upsample the in-phasecomponent to provide an upsampled in-phase signal; and a digital lowpass filter configured to filter the upsampled in-phase signal toprovide the processed in-phase signal, and wherein the second digitalprocessing branch comprises: a second upsampler configured to upsamplethe quadrature phase component to provide an upsampled quadrature phasesignal; and a second digital low pass filter configured to filter theupsampled quadrature phase signal to provide the processed quadraturephase signal.
 33. The transmitter of claim 31, wherein the first analogprocessing branch comprises: a first digital to analog converterconfigured to convert the amplitude compressed in-phase signal from adigital domain to an analog domain to provide an analog amplitudecompressed in-phase signal; a first analog low pass filter configured tofilter the analog amplitude compressed in-phase signal to provide theprocessed amplitude compressed in-phase signal, and wherein the secondanalog processing branch comprises: a second digital to analog converterconfigured to convert the amplitude compressed quadrature phase signalfrom the digital domain to the analog domain to provide an analogamplitude compressed quadrature phase signal; a second analog low passfilter configured to filter the analog amplitude compressed quadraturephase signal to provide the processed amplitude compressed quadraturephase signal.
 34. The transmitter of claim 31, wherein the compressivenonlinearity module is further configured to amplitude compress theprocessed in-phase signal and the processed quadrature phase signalusing a first and a second compressive nonlinearity function,respectively.
 35. The transmitter of claim 34, wherein compressions ofthe first and the second compressive nonlinearity functions increasewith an amplitude of the processed in-phase signal and an amplitude ofthe processed quadrature phase signal, respectively.
 36. The transmitterof claim 34, wherein the first and the second compressive nonlinearityfunctions are configured to enable compliance with a spectral mask asmeasured at an output of the amplifier.
 37. The transmitter of claim 34,wherein the first and the second compressive nonlinearity functions areconfigured to substantially preserve an angle of the in-phase andquadrature phase components.
 38. A transmitter, comprising: an inverseFast-Fourier transform (IFFT) module configured to implement an IFFT onsymbol data to provide a time-domain representation of the symbol data;a compressive nonlinearity module configured to provide an amplitudecompressed time-domain representation in response to the time-domainrepresentation; an amplifier configured to provide an amplified outputhaving a power level that is greater than a rated power level of theamplifier module in response to the amplitude compressed time-domainrepresentation.
 39. The transmitter of claim 38, wherein the compressivenonlinearity module is further configured to amplitude compress itsinput using a compressive nonlinearity function.
 40. The transmitter ofclaim 39, wherein compression of the compressive nonlinearity functionincreases with an amplitude of its input.
 41. The transmitter of claim39, wherein the compressive nonlinearity function is configured toenable compliance with a spectral mask as measured at an output of theamplifier.
 42. The transmitter of claim 38, further comprising: anupsampler configured to upsample the time-domain representation toprovide an unsampled time-domain representation; and a digital low passfilter configured to filter the upsampled time-domain representation toprovide a filtered time-domain representation, wherein the compressivenonlinearity module is further configured to amplitude compress thefiltered time-domain representation to provide the amplitude compressedtime-domain representation.
 43. The transmitter of claim 38, furthercomprising: a digital to analog converter configured to convert theamplitude compressed time-domain representation from a digital domain toan analog domain to provide an analog amplitude compressed time-domainrepresentation; an analog low pass filter configured to filter theanalog amplitude compressed time-domain representation to provide afiltered amplitude compressed time-domain representation; and anupconverter module configured to frequency translate the filteredamplitude compressed time-domain representation to provide a frequencytranslated amplitude compressed time-domain representation, wherein theamplifier is configured to amplify the frequency translated amplitudecompressed time-domain representation to provide the amplified output.44. A transmitter, comprising: a compressive nonlinearity moduleconfigured to amplitude compress a time-domain representation of symboldata in a time-domain using a compressive nonlinearity function toprovide an amplitude compressed time-domain representation; and anamplifier configured to provide an amplified output having a power levelthat is greater than a rated power level of the amplifier module inresponse to the amplitude compressed time-domain representation.
 45. Thetransmitter of claim 44, wherein compression of the compressivenonlinearity function increases with an amplitude of the time-domainrepresentation of the symbol data.
 46. The transmitter of claim 44,wherein the compressive nonlinearity function is configured to enablecompliance with a spectral mask as measured at an output of theamplifier.
 47. The transmitter of claim 44, wherein the symbol data isOrthogonal Frequency Division Multiplexed (OFDM) symbol data.
 48. Thetransmitter of claim 47, further comprising: an inverse Fast-Fouriertransform (IFFT) module configured to implement an IFFT on the OFDMsymbol data to provide a time-domain representation of the OFDM symboldata; a digital processing branch configured to process the time-domainrepresentation to provide a processed time-domain representation;wherein the compressive nonlinearity module is further configured toamplitude compress the processed time-domain representation.
 49. Thetransmitter of claim 48, wherein the digital processing branchcomprises: an upsampler configured to upsample the time-domainrepresentation to provide an upsampled time-domain representation; and adigital low pass filter configured to filter the upsampled time-domainrepresentation to provide the processed time-domain representation. 50.The transmitter of claim 47, further comprising: an analog processingbranch configured to process an output of the compressive nonlinearitymodule to provide an analog processed time-domain representation; anupconverter module configured to frequency translate the analogprocessed time-domain representation to provide a frequency translatedtime-domain representation, wherein the amplifier is configured toamplify the frequency translated time-domain representation to providethe amplified output.
 51. The transmitter of claim 50, wherein theanalog processing branch comprises: a digital to analog converterconfigured to convert the output of the compressive nonlinearity modulefrom a digital domain to an analog domain to provide an analog amplitudecompressed time-domain representation; and an analog low pass filterconfigured to filter the analog amplitude compressed time-domainrepresentation to provide the analog processed time-domainrepresentation.